Semiconductor memory device

ABSTRACT

A memory device includes an error detection and correction system with an error correcting code over GF(2 n ), wherein the system has an operation circuit configured to execute addition/subtraction with modulo 2 n −1, and wherein the operation circuit has a first operation part for performing addition/subtraction with modulo M and a second operation part for performing addition/subtraction with modulo N (where, M and N are integers which are prime with each other as being obtained by factorizing 2 n −1), and wherein the first and second operation parts perform addition/subtraction in parallel to output an operation result of the addition/subtraction with modulo 2 n −1.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of application No. 11/674,384, filed Feb. 13, 2007, and is based on and claims the benefit of priority from the prior Japanese Patent Application No. 2006-042250, filed on Feb. 20, 2006, the entire contents of which are incorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a semiconductor memory device and, more particularly, to an on-chip error detection and correction system adaptable for use therein.

2. Description of the Related Art

Electrically rewritable nonvolatile semiconductor memory devices, i.e., flash memories, increase in error rate with an increase in number of data rewrite operations. In particular, the quest for larger storage capacity and further enhanced miniaturization results in an increase in error rate. In view of this, an attempt is made to mount or “embed” a built-in error correcting code (ECC) circuit on flash memory chips or, alternatively, in memory controllers for control of these chips. An exemplary device using this technique is disclosed, for example, in JP-A-2000-173289.

A host device side using more than one flash memory is designable to have an ECC system which detects and corrects errors occurring in the flash memory. In this case, however, the host device increases in its workload when the error rate increases. For example, it is known that a two-bit error correctable ECC system becomes greater in calculation scale, as suggested by JP-A-2004-152300.

Accordingly, in order to cope with such error rate increase while suppressing the load increase of the host device, it is desired to build a 2-bit error correctable ECC system in the flash memory. What is needed in this case is to meet the conflicting requirements: i.e., increasing the ECC system's arithmetic operation speed, and yet lessening possible penalties of read/write speed reduction of the flash memory.

SUMMARY OF THE INVENTION

According to one aspect of the present invention, there is provided a semiconductor memory device including an error detection and correction system with an error correcting code over Galois field GF(2^(n)), wherein

the error detection and correction system includes an operation circuit configured to execute addition/subtraction with modulo 2^(n)−1, and wherein the operation circuit includes a first operation part for performing addition/subtraction with modulo M and a second operation part for performing addition/subtraction with modulo N (where, M and N are integers, which are prime with each other as being obtained by factorizing 2^(n)−1), which perform addition/subtraction simultaneously in parallel with each other to output an operation result of the addition/subtraction with modulo 2^(n)−1.

According to another aspect of the present invention, there is provided a semiconductor memory device including a cell array with electrically rewritable and non-volatile memory cells arranged therein, and an error detection and correction system, which is correctable up to 2-bit errors for read out data of the cell array by use of a BCH code over Galois field GF(256), wherein

the error detection and correction system includes:

an encoding part configured to generate check bits to be written into the cell array together with to-be-written data;

a syndrome operating part configured to execute syndrome operation for read out data of the cell array;

an error location searching part configured to search error locations in the read out data based on the operation result of the syndrome operating part; and

an error correcting part configured to invert an error bit in the read out data detected in the error location searching part, and output it, and wherein

the error location searching part includes an operation circuit configured to execute index addition/subtraction with modulo 255, and wherein

the operation circuit includes a first operation part for performing addition/subtraction with modulo 17 and a second operation part for performing addition/subtraction with modulo 15, which perform addition/subtraction simultaneously in parallel with each other to output an operation result of the index addition/subtraction with modulo 255.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram showing a configuration of main part of a flash memory in accordance with an embodiment of this invention.

FIG. 2 is a diagram showing a detailed arrangement of a cell array in the flash memory.

FIG. 3 is a diagram showing a further detailed configuration of the cell array.

FIG. 4 is a diagram showing a data level relationship of the flash memory.

FIG. 5A is a diagram showing a 4-bit parity check circuit as used in an ECC circuit for performing even/odd N judgment of “1”; and FIG. 5B is a diagram showing symbols of the parity checker circuit.

FIG. 6 is a diagram showing a product computation method of polynomials over GF(2).

FIG. 7 is a diagram showing a parity check circuit for use in such product computation.

FIG. 8 is a diagram showing a check bit calculation method in the encoding part in the ECC circuit.

FIG. 9 is a diagram showing a calculation method of a syndrome polynomial S₁(x) in the decoding part in the ECC circuit.

FIG. 10 is a diagram showing a calculation method of syndrome polynomial S₃(x³) in the same.

FIG. 11 is a diagram showing 144 degrees as selected from an information polynomial in order to be used as data w bits.

FIG. 12 is a table of “n”s with the coefficient of each degree of the 15-degree remainder polynomial being of “1”.

FIG. 13 is a diagram showing a configuration of a parity check circuit for use in check bit calculation.

FIG. 14 is a table of n's with the coefficient of each order of “1” at the selected n's of remainder polynomial p^(n)(x) as used in calculation of the syndrome polynomial S₁(x).

FIG. 15 shows an exemplary circuit for calculation of the syndrome polynomial S₁(x).

FIG. 16 is a table of n's with each degree coefficient “1” at chosen n's of a remainder polynomial p^(3n)(x) for use in calculation of a syndrome polynomial S₃(x³).

FIG. 17 shows an exemplary circuit for calculation of the syndrome polynomial S₃(x³).

FIG. 18 is a coefficient table of remainder polynomial p^(n)(x) as need in decoding.

FIGS. 19A and 19B are tables for showing the relationships between indexes n and y_(n).

FIG. 20 shows the summarized relationships between n and y_(n).

FIG. 21 is a table showing classification of index y_(n) based on 15 y _(n)(17) and 17 y _(n)(15).

FIG. 22 is a table showing the relationships between index i and the corresponding physical data bit position k.

FIG. 23A shows an index rotator for performing addition/subtraction of indexes; and FIG. 23B circuit symbol thereof.

FIG. 24 shows the index rotator part for operating the first congruence in Expression 18.

FIG. 25 shows the decode circuit used in FIG. 25.

FIG. 26 is a table showing index n as the remainder class index 15 n(17).

FIG. 27 is a table showing index n as the remainder class index −45 n(17).

FIG. 28 shows the index rotator part for operating the second congruence in Expression 18.

FIG. 29 is a table showing index n as the remainder class index 17 n(15).

FIG. 30 is a table showing index n as the remainder class index −17 n(15).

FIG. 31 shows output buses for integrating outputs of the index rotators shown in FIGS. 24 and 28.

FIG. 32 shows the index rotator part for operating the first congruence in Expression 19.

FIG. 33 shows the decode circuit used in FIG. 32.

FIG. 34 is a table showing the relationships between 15 y _(n)(17), 17 y _(n)(15) and 15 n(17).

FIG. 35 shows the index rotator part for operating the second congruence in Expression 19.

FIG. 36 is a code table of the decode circuit in FIG. 35.

FIG. 37 shows error correction part for integrating the outputs of index rotators shown in FIGS. 32 and 35 to output error bit location signals.

FIG. 38 is a table showing the relationships between 15 i(17), 17 i(15), i and k.

FIG. 39 shows the error correction circuit.

FIG. 40 shows the 2-bit parity check circuit used in FIG. 39.

FIG. 41 is a diagram showing a configuration of the ECC circuit relating to one block memory core.

FIG. 42 shows another memory core configuration.

FIG. 43 shows another embodiment applied to a digital still camera.

FIG. 44 shows an internal configuration of the digital still camera.

FIGS. 45A to 45J show other electric devices to which the embodiment is applied.

DETAILED DESCRIPTION OF THE EMBODIMENTS

Some embodiments of this invention will be described with reference to the accompanying figures of the drawing below.

Referring to FIG. 1, there is illustrated a block diagram showing main parts of a flash memory embodying the invention. This flash memory includes a couple of banks BNK0 and BNK1. Bank BNK0 is formed of four separate memory cell arrays 1—i.e., T-cell array (0-1), C-cell array (0-2), T-cell array (0-3), and C-cell array (0-4). The other bank BNK1 also is formed of four cell arrays 1, i.e., T-cell array (1-1), C-cell array (1-2), T-cell array (1-3), and C-cell array (1-4).

Each cell array 1 is associated with a row decoder (RDEC) 3 for performing word-line selection. Sense amplifier (SA) circuits 2 are provided, each of which is commonly owned or “shared” by neighboring T-cell array and C-cell array. These T-cell and C-cell arrays have a plurality of information cells, T-cells and C-cells, respectively, each having at least one reference cell R-cell as will be described later in detail.

The information cells T-cell and C-cell are the same in structure as the reference cell R-cell. Upon selection of an information cell T-cell from T-cell array, a reference cell R-cell is selected from a C-cell array that makes a pair with this T-cell array. Similarly, when an information cell C-cell is selected from C-cell array, a reference cell R-cell is selected from a T-cell array that makes a pair therewith.

Read/write data of upper and lower cell array groups each having four separate cell arrays 1 are transferred between sense amplifier circuits 2 and external input/output (I/O) nodes through data buses 5 and 6 via an I/O buffer 7. Between the upper and lower cell array groups each having four cell arrays 1, an error correcting code (ECC) circuit 8 is provided as an error detection and correction system, which is operable to detect and correct errors of read data.

Referring next to FIG. 2, a detailed configuration of a pair of T-cell array and C-cell array with their shared sense amp circuit 2 is shown. T-cell array has parallel extending bit lines BL, each of which is connected to a plurality of information cell NAND strings, T-NAND, and at least one reference cell NAND string, R-NAND. Similarly C-cell array has parallel bit lines BBL, to each of which are connected a plurality of information cell NAND strings, C-NAND, and at least one reference cell NAND string, R-NAND. Bit line BL of T-cell array and its corresponding bit line BBL in C-cell array constitute a pair.

The sense amp circuit 2 has sense units SAU of a current detection type, each of which is for detecting a difference between currents flowing in a pair of bitlines BL and BBL to sense data. Although in FIG. 2 a sense unit SAU is disposed one by one between paired bit lines BL and BBL, there is usually employed a scheme for causing a m single sense unit to be shared by more than two bit line pairs.

A prespecified number of serial combinations of the information cell NAND strings T-NAND, C-NAND and reference cell NAND string R-NAND are laid out in the direction at right angles to the bitlines, thereby constituting cell blocks. In the respective cell blocks, word lines TWL, CWL and RWL are disposed. More specifically, these cell blocks each has a plurality of groups of parallel word lines which insulatively cross or “intersect” the bit lines BL and BBL—i.e., bundles of word lines TWL associated with respective columns of information cell NAND strings T-NAND, sets of wordlines CWL connected to respective columns of information cell NAND strings C-NAND, and wordlines coupled to each column of reference cell NAND strings R-NAND.

FIG. 3 shows a detailed configuration of circuitry which includes a sense unit SAU and an information cell NAND string T-NAND (or C-NAND) and a reference cell NAND string R-NAND as connected to the sense unit. Each of these NAND strings has a serial connection of electrically rewritable nonvolatile memory cells M0 to M31 and a couple of select transistors SG1 and SG2 at opposite ends thereof. While the nonvolatile memory cells M0-M31 used are the same in transistor structure, these may function as the information cells T-cell (or C-cell) in information cell NAND string T-NAND (or C-NAND) or act as the reference cell R-cell in reference cell NAND string R-NAND.

In a data sense event, a memory cell of the information cell NAND string T-NAND (or C-NAND) and its corresponding cell in reference cell NAND string R-NAND are selected together at a time. This simultaneous data/reference cell selection results in currents Ic and Ir flowing in these cell strings, respectively. The sense unit SAU detects a difference between these cell currents Ic and Ir to sense data.

FIG. 4 shows a distribution of memory cell data levels (threshold levels) in case of four-level storage scheme. Any one of four different data levels L0, L1, L2 and L3 is written into an information cell T-cell, C-cell. A reference level Lr is written into reference cell R-cell, which level is set to have a potential between the data levels L0 and L1 for example.

Bit allocation of four data levels L0-L3 is different between the information cells T-cell and C-cell. For example, suppose that four-level data is represented by (HB, LB), where HB is the upper-level bit; and LB lower-level bit. In an information cell T-cell on the T-cell array side, data bits are assigned as follows: L0=(1,0), L1=(1,1), L2=(0,1), and L3=(0,0). While, in an information cell C-cell on the C-cell array side, data bits are assigned as follows: L0=(0,0), L1=(0,1), L2=(1,1), and L3=(1,0).

In FIG. 4, there are shown read voltages R1 to R3 given to information cell T-cell, C-cell during reading in a way pursuant to the data to be read, and a read voltage Rr applied to reference cell R-cell. Also shown here are write-verify voltages P1 to P3 to be given to the information cell T-cell, C-cell during write-verifying, and a write-verify voltage Pr applied to the reference cell R-cell.

Although, in the example, four-level data storage scheme has been explained, it will be used in general such a multi-level data storage scheme that two or more bits are stored in each memory cell.

An explanation will now be given of a technique used in this embodiment for mounting the ECC circuit 8 and its access mode in a case where a 1 gigabit (Gb) of memory is configured using two 512-megabit (Mb) banks BNK0 and BNK1.

The memory as discussed here is of a x16IO configuration, in which address generation is in common to all the banks although each bank's page address is settable independently and wherein allocation is made to each bank by designating which page address is applied to which bank. Accordingly, interleaving is done between the banks for page address utilization.

Each bank has 1,024 k pages. The page length defined in common for 16 IOs of each bank is 32 bits, which may be output 8 bits by 8 bits in a parallel way. The page length is the maximum or “longest” data length per bank with respect to one IO readable by a sense operation with one-time wordline address setting.

With such an arrangement, 128 bits of data are output outside by one-time data transmission from the sense amp circuit 2. In other words, this design permits 2^(k) data bits to be transferred together, where k is an integer.

The ECC circuit 8 built in the flash memory is arranged to employ Bose-Chaudhuri-Hocquenghem (BCH) code to with 2-bit error correctability—that is, double error correction BCH code system, which will be referred to as “2EC-BCH” code hereinafter. To enable 2-bit error correction, a need is felt to use a simultaneous equation having different roots. The 2EC-BCH code is a cyclic code generated with a code generation polynomial as denoted by a product of two primitive polynomials.

The bit length used here is given by 2^(n)−1. Data bits usable as information are 2^(n)−1−2n bits. To deal with 2^(m) bits of data, let n=m+1. This makes it inevitable to use a 2o data length that is approximately two times greater than the quantity required.

In the memory configuration of FIG. 1, 2EC-BCH code used for execution of 2-bit error correction in 128-bit information data is one over Galois field GF(2⁸). In this case, usable bit length is 2⁸−1=255, which requires the use of 16 bits as error check bits. Thus, in case 144 bits are used for 16 check bits and 128 information data bits, the remaining 111 bits become extra ones. In this case, as shown in FIG. 1, the data buses 5 extending from upper and lower cells which bisect each bank are 72 DQ lines, respectively, resulting in a total of 144 bits of data being transferred together at a time.

The ECC system is variable in efficiency depending upon the handling of useless extra 111 bits and how information bit selection is performed in the BCH code system. Thus, it is necessary to take into consideration a method for configuring the best suited ECC system.

Although in some cases the required bit number becomes greater than 128 bits when considered also including the redundancy for replacement of a defective cell(s), this may readily be analogously extendable from the case of the 128-bit data length discussed here. In general, the number L of data bits to be error-corrected in the 2EC-BCH system is selected in the range of L≦255−16=239.

(Data Encoding)

An explanation will be given of the outline of 2EC-BCH over Galois field GF(2⁸). Letting a primitive root (element) of GF(256) be α, 8-degree primitive polynomial m₁ (x) on the ground field GF(2) with this element α being as its own root is represented by Expression 1. In other words, irreducible polynomials of a power of α and a power of x due to m₁(x) become mutually corresponding elements in GF(256).

Additionally, as an 8-degree irreducible polynomial with a cubic of a being as its root, polynomial m₃(x) that is relatively prime with m₁(x) is used as shown in the following Expression 1.

α: m ₁(x)=x ⁸ +x ⁴ +x ³ +x ²+1

α³ : m ₃(x)=x ⁸ +x ⁶ +x ⁵ +x ⁴ +x ² +x+1   [Exp. 1]

Based on these two primitive polynomials, a 2-bit error correctable ECC system (i.e., 2EC-BCH code system) will be configured. To perform encoding with check bits added to the data being written, prepare a product polynomial g(x) of m₁(x) and m₃(x) as a code generation polynomial, as shown in Expression 2 below.

$\begin{matrix} \begin{matrix} {{g(x)} = {{m_{1}(x)}{m_{3}(x)}}} \\ {= {x^{16} + x^{14} + x^{13} + x^{11} + x^{10} + x^{9} + x^{8} + x^{6} + x^{5} + x + 1}} \end{matrix} & \left\lbrack {{Exp}.\mspace{14mu} 2} \right\rbrack \end{matrix}$

A maximal number of two-bit error correctable bits capable of being utilized as information bits is 239, which is obtained by subtracting check bit numbers, 16, from 2⁸−1 (=255). Based on these bits, while letting coefficients from bit positions 16 to 254 be a₁₆ to a₂₅₄, make a 238-degree information polynomial f(x) as indicated by Expression 3.

f(x)=a ₂₅₄ x ²³⁸ +a ₂₅₃ x ²³⁷ + . . . +a ₁₈ x ² +a ₁₇ x+a ₁₆   [Exp. 3]

For example, 128 bits are actually used in the 239 terms. In this case, letting the remaining coefficients of 111 bits be fixed to “0”, the information polynomial becomes one with the lack of those terms of corresponding is degrees. Depending upon which degree numbers are selected as the 111 terms with such “0”-fixed coefficients from the information polynomial f(x) having 239 terms, the computation amount of syndrome calculation becomes different, which is to be executed during decoding as described later. Thus, this selection technique becomes important. This will be explained later.

From the information polynomial f(x), form a data polynomial f(x)x¹⁶ that contains 16 check bits. To make such check bits from this data polynomial, the data polynomial f(x)x¹⁶ will be divided by the code generation polynomial g(x) to obtain 15-degree remainder polynomial r(x) as shown in the following Expression 4.

f(x)x ¹⁶ =q(x)g(x)+r(x)

r(x)=b ₁₅ x ¹⁵ +b ₁₄ x ¹⁴ + . . . +b ₁ x+b ₀   [Exp. 4]

Use the coefficients b₁₅ to b₀ of this remainder polynomial r(x) as the check bits. In other words, 128 coefficients a_(i(128)) to a_(i(1)) selected from 239 ones serve as “information bits” while 16 bits of b₁₅ to b₀ serve as “check bits”, thereby resulting in that a total of 144 bits become “data bits” to be stored in the memory as shown in the following Expression 5.

a_(i(128))a_(i(127)) . . . a_(i(3))a_(i(2))a_(i(1))b₁₅b₁₄ . . . b₁b₀   [Exp. 5]

Here, a_(i(k)) is the data to be externally written into the memory. Based on this data, a check bit b_(j) is created in the chip-embedded ECC system, which bit will be simultaneously written into the cell array.

(Data Decoding)

Next, an explanation will be given of a method for detecting errors from 144 bits of data read out of the cell array and for correcting up to 2 bits of errors.

Supposing that errors take place when the memory stores the coefficients of 254-degree data polynomial f(x)x¹⁶, such errors also are expressed by 254-degree polynomial. This error polynomial being e(x), the data read from the memory may be given by a polynomial v(x) with a structure shown in Expression 6 as follows.

v(x)=f(x)x ¹⁶ +r(x)+e(x)   [Exp. 6]

A term with the coefficient of this error polynomial e(x) in Expression 6 being at “1” is identical to an error. In other words, detecting e(x) is equivalent to performing error detection and correction.

What is to be done first is to divide the readout data polynomial v(x) by the primitive polynomials m₁(x) and m₃(x) to obtain the respective remainders, which are given as S₁(x), S₃(x). As shown in Expression 7, it is apparent from the structure of v(x) that each is equal to the remainder of e (x) divided by m₁(x), m₃ (x).

v(x)≡S₁(x)mod m₁(x)→e(x)≡S₁(x)mod m₁(x)

v(x)≡S₃(x)mod m₃(x)→e(x)≡S₃(x)mod m₃(x)   [Exp. 7]

These division remainders S₁(x) and S₃(x) are referred to as syndrome polynomials.

Assuming that 2-bit errors are present at i-th and j-th bits, e(x) will be expressed as follows: e(x)=x^(i)+x^(j). These values i and j are obtainable by calculation of the index “n” of x=α^(n), i.e., a root of m₁(x) that is an element in GF(256). More specifically, when letting a remainder, w which is obtained by dividing x^(n) by m₁(x), be p^(n)(x), α^(n)=p^(n)(x). As shown in the following Expression 8, let α^(i) and α^(j) corresponding to error degrees be X₁ and X₂, respectively; let the indexes corresponding to S₁(α) and S₃(α³) with respect to syndromes S₁ (x) and S₃(x) be σ₁ and σ₃; and let S₁(α) and S₃(α³) be S₁ and S₃, respectively.

X ₁ =p ^(i)(α)=α^(i)

X ₂ =p ^(j)(α)=α^(j)

S ₁(α)=S ₁=α^(σ1)

S ₃(α³)=S ₃=α^(σ3)   [Exp. 8]

Since m₃(α³)=0, we obtain the following Expression 9.

S ₁ =X ₁ +X ₂ =e(α)

S ₃ =X ₁ ³ +X ₂ ³ =e(α³)   [Exp. 9]

At the second stage, consider polynomial Λ^(R)(x) with unknown quantities X₁ and X₂ as its roots, product X₁X₂ is representable by S₁ and S₃ as in Expression 10, so that the coefficients involved are calculable from the syndrome polynomials.

$\begin{matrix} \begin{matrix} {{S_{3}/S_{1}} = {\left( {X_{1}^{3} + X_{2}^{3}} \right)/\left( {X_{1} + X_{2}} \right)}} \\ {= {X_{1}^{2} + {X_{1}X_{2}} + X_{2}^{2}}} \\ {= {\left( {X_{1} + X_{2}} \right)^{2} + {X_{1}X_{2}}}} \\ {= {S_{1}^{2} + {X_{1}X_{2}}}} \\ {{X_{1}X_{2}} = {\left( {S_{3} + S_{1}^{3}} \right)/S_{1}}} \\ {{\Lambda^{R}(x)} = {\left( {x - X_{1}} \right)\left( {x - X_{2}} \right)}} \\ {= {x^{2} + {S_{1}x} + {\left( {S_{3} + S_{1}^{3}} \right)/S_{1}}}} \\ {= {x^{2} + {\alpha^{\sigma 1}x} + \alpha^{{\sigma 3} - {\sigma 1}} + \alpha^{2{\sigma 1}}}} \end{matrix} & \left\lbrack {{Exp}.\mspace{14mu} 10} \right\rbrack \end{matrix}$

At the third stage, finding α^(n), i.e., a root of Λ^(R)(x) in GF(256), it becomes possible to obtain the error bit locations i and j as “n” of α^(n) from X₁, X₂=α^(n). In other words, searching Λ^(R)(α^(n))=0 for n=0, 1, 2, . . . , 254, a hit number “n” will be specified as an error bit.

As shown in Expression 11 below, in case of a 1-bit error, we obtain X₁=S₁, X₁ ³=S₃=S₁ ³. Thus, the error location is defined from S₁. If there are no errors, we obtain S₁=S₃=0. In case there are 3 bits or more errors and its position is incomputable, either one of S₁ and S₃ becomes 0.

(a) If 1-bit error, X ₁ =S ₁ and X ₁ ³ =S ₃ =S ₁ ³.

(b) If 0-bit error, S ₁ =S ₃=0.

(c) If more than 3-bit error, S₁ or S₃ is equal to 0.   [Exp. 11]

As described above, error location searching is for obtaining index “n” of α^(n) that satisfies Λ^(R)(x)=0. For the purpose, in this embodiment, change Λ^(R)(x) shown in Expression 10, and make possible to obtain “n” by use of only index relationships. In detail, using the conversion of: x=α^(σ1)y, to solve Λ^(R)(x)=0, and to obtain variable y shown in the following Expression 12, it becomes equal to each other.

y ² +y+1+α^(σ3−3σ1)=0   [Exp. 12]

By use of this Expression 12, directly comparing the index obtained by variable calculation with that defined by syndrome calculation, it is possible to find a coincident variable. In detail, to solve the Expression 12, substitute α^(n) for y to obtain the index y_(n) shown in Expression 13.

y ² +y+1=α^(2n)+α+1=α^(yn)   [Exp. 13]

As shown in the following Expression 14, comparing the index σ₃−3σ₁ obtained by the syndrome calculation with that y_(n) obtained by the variable calculation, coincident “n” becomes the index of y corresponding to the error location.

σ₃−3σ₁≡y_(n) mod 255   [Exp. 14]

To restore the index of variable y to that of the real variable x, as shown in Expression 15, multiply α^(σ1) into y.

x=α ^(σ1) y=α ^(σ1+n)   [Exp. 15]

The index σ₁+n of α as shown in Expression 15 is that of x corresponding to the error location, and this x will satisfy the equation of: Λ^(R)(x)=0.

Essentials for execution of actual calculations are summarized as follows.

What is needed in encoding is a remainder table, i.e., a table of coefficients of remainder polynomial r(x), which is generated by code generation polynomial g(x) from 128 terms selected as data bits from the data polynomial f(x)x¹⁶ of degree 254 in maximum. For check bit calculation, select those coefficients corresponding to the data bit-selected terms, followed by execution of addition over GF(2) using two-element code of “0” or “1”.

In decoding, when performing calculation of the syndrome polynomials S₁(x) and S₃(x³), a remainder table is necessary, which is a table of coefficients of remainder p^(n)(x) obtained by primitive polynomial m₁(x) from 254 to 0 degree. Based on this table, the calculation is done as similar to the check bit calculation.

To shorten an ECC calculation time in the memory system which does not use all of the 239 data bits usable in 2EC-BCH using GF(256), it is in need of employing a practical selection method for selecting actually used s terms (degrees) from the information polynomial. Especially for the syndrome calculation, it is necessary to choose specific terms (degrees) capable of efficiently obtaining the remainder.

An explanation will first be given of a practical remainder calculation method.

In calculations over GF(2), both multiplication and division are carried out by addition of polynomial coefficients—that is, based on even/odd judgment of the numbers of “1”. Thus a calculator circuit used here is principally designed to perform parity check in a way as follows: if the number of terms, coefficient of which is “1”, is an even number, output a computation result “0”; if it is an odd number then its calculation result becomes “1”.

A 4-bit parity check (PC) circuit with a simplified configuration is shown in FIG. 5A, a circuit symbol of which is shown in FIG. 5B. While parity checkable circuitry with various configurations is designable and modifiable on a case-by-case basis, the illustrative circuit is arranged to have parity-check logic units made up of transistors only for purposes of convenience in illustration and discussion herein.

This parity check circuit includes an upper-stage circuit 401 which receives at its inputs four bits a, b, c, d and their complementary signals /a, /b, /c, /d and generates “0” at an output node EP exclusively when an even number of “1”s are in the input signals, and a lower-stage circuit 402 which generates “0” at its output node OP only when an odd number of “1”s are in the input signals thereof. These circuits 401-402 are formed of eight parallel-connected gates between the power supply voltage Vdd and ground potential Vss, each of which gates has four 4-bit input PMOS transistors and NMOS transistors.

More specifically, the upper-stage circuit 401 is such that when its “1” inputs are 0, 2 or 4 in number, NMOS transistors on the Vss side are rendered conductive, causing the EP's potential to be “0” (EP=“0”). For the lower-stage circuit 402, when the number of its inputs “1”s is 1 or 3, the Vss-side NMOS transistors turn on, resulting in establishment of OP=“0”.

With this parity check circuit, 4-bit parity check is calculable within a shortened delay time, which is equivalent to that of a single stage of inverter.

A calculation method of a product over GF(2) of the polynomial in GF(256) using the above-described parity is check circuit is shown in FIG. 6. Every arithmetic processing of the polynomial in GF(256) becomes computation between 7-degree polynomials p^(n)(x) because the primitive polynomial is of degree 8. Thus, every calculation result of four basic operations—i.e., addition, subtraction, multiplication and division—becomes any one of the polynomials p^(n)(x). Division {p^(n)(x)}⁻¹ becomes a product of p^(255−n)(x).

While letting the coefficient of m-degree term of p^(n)(x) be represented by P^(n) _(m), a product of 7-degree polynomials p^(i)(x) and p^(j)(x) in Expression 16 below is calculated as shown in FIG. 6. Obviously, the coefficient P^(n) _(m) is either “0” or “1”.

p ^(i)(x)=P ^(i) ₇ x ⁷ +P ^(i) ₆ x ⁶ +P ^(i) ₅ x ⁵ +P ^(i) ₄ x ⁴ +P ^(i) ₃ x ³ +P ^(i) ₂ x ² +P ^(i) ₁ x+P ^(i) ₀

p ^(j)(x)=P ^(j) ₇ x ⁷ +P ^(j) ₆ x ⁶ +P ^(j) ₅ x ⁵ +P ^(j) ₄ x ⁴ +P ^(j) ₃ x ³ +P ^(j) ₂ x ² +P ^(j) ₁ x+P ^(j) ₀   [Exp. 16]

As x^(n)≡p^(n)(x) (mod m₁(x)), use such a rule that the 14-degree term is with p^(j+7) appearing 7-item ahead of the multiplying p^(j)(x). Then, the product of polynomials p^(i)(x) and p^(j)(x) is enabled by multiplication and addition (i.e., parity check) between the respective polynomial coefficients as shown in FIG. 6.

In other words, parity check is done which is per-degree addition of from 7 to 0 of 1- or 0-multiplied ones, and its result becomes the coefficient of each degree of p^(i)(x)p^(j)(x). Regarding the multiplying pj(x), there are required those polynomial coefficients up to a 7-degree ahead one.

The circuitry required here may be the parity check w circuit only, with additional provision of a circuit for performing inputting unique to the coefficients of from the multiplying p^(j)(x) to p^(j+7)(x). This is because no parity check is needed when the coefficient P^(j) _(m)=0. As shown in FIG. 7, the parity check circuit is modifiable to employ 3-bit parity check circuit, 2-bit parity check circuit other than the 4-bit parity check circuit in accordance with input number needed.

This computation method is applied upon generation of the check bits during encoding, which will be explained below.

The check bit calculation is performed as follows: dividing data polynomial f(x)x¹⁶ generated from information data by code generation polynomial g(x), thereby obtaining remainder polynomial r(x). To perform this arithmetic operation, precalculate coefficients of 15-degree remainder polynomial r^(i)(x) which was obtained by dividing a single term x^(i) by g(x), and use them in a similar way to the case of the multiplication of p^(n)(x). The coefficient of r^(i)(x) is referred to as R^(i) _(m)(m=0, 1, 2, . . . , 15), and the coefficient of x^(i) of f(x)x¹⁶, which serves as an information bit, is referred to as a_(i).

Actual calculation is as follows. Assuming that data bits are a₁₆ to a₂₅₄, 111 bits out of them are selected in a way such that the syndrome calculation to be done later becomes less, and then fixed to “0”. Next, as shown in FIG. 8, perform multiplication of the data polynomial's coefficient a_(i) and the remainder polynomial's coefficient R^(i) _(m) and addition (parity check) of those coefficients of terms of the same degree.

More specifically, perform addition—i.e., parity s check—of the coefficient R^(i) _(m) of a remainder r^(i)(x) of specific degree with the data bit a_(i) being at “1” in GF(2) per each m; then, let the result be a check bit b_(m).

An explanation will next be given of a calculation method during decoding as required when performing the selection of “0”-fixed bit positions of 111 bits.

Firstly, the calculation of syndrome polynomial S₁(x) is done, which is for obtaining a division remainder at m₁(x) of polynomial v(x) with data d_(i) as read out of a memory cell being as its coefficient. This arithmetic is operation is performed, as shown in FIG. 9, by multiplication of d_(i) and the coefficient P^(i) _(m) (m is 0, 1, . . . , 7) of the remainder p^(i)(x) that is obtained by dividing x^(i) (i is 254, 253, . . . , 0) by m₁(x), and addition (parity check) of its result. In other words, let a parity check result of m-th degree coefficient P^(i) _(m) of p^(i)(x) with d_(i)=“1” be the m-degree coefficient of the syndrome polynomial S₁(x).

In this calculation procedure, no calculations are done for portions of d_(i)=“0” and when P^(i) _(m) is “0”, so that the selection of out-of-use bits determines the calculation amount in the case where all of 239 information bits are not used.

A calculation method relating to another syndrome polynomial S₃(x) will next be described below.

In relation to the syndrome polynomial S₃(x), what is m needed for searching the error location j is the syndrome polynomial S₃(x³). Note that S₃(x) per se is a division remainder at m₃(x) of v(x). Between v(x³) and S₃(x³), a relationship shown in the following Expression 17 is establishable, where P^(i) _(m)(m=0, 1, . . . , 7) is the coefficient of remainder polynomial p^(i)(x) obtained by dividing x^(i) by m_(i)(x), and d_(i) is the coefficient of x^(i).

S₃(x)≡v(x)mod m₃(x),

m₃(x³)≡0 mod m₁(x)

S₃(x³)≡v(x³)mod m₃(x³)≡v(x³)mod m₁(x)

v(x)≡Σd_(i)x^(i),

v(x³)≡Σd_(i)x^(3i),

x^(i)≡p^(i)(x)mod m₁(x)

x^(3i)≡p^(3i)(x)mod m₁(x)

S₃(x³)≡v(x³)mod m₁(x)≡Σd_(i)p^(3i)(x)mod m₁(x).   [Exp. 17]

From the above-described Expression 17, S₃(x³) is calculable by the coefficient d_(i) of v(x) and the remainder p^(3i)(x). Thus, what is needed here is the coefficient P^(i) _(m) of p^(i)(x) at m₁(x) of x^(i), and its practical calculation method is as shown in FIG. 10.

In this calculation process also, no calculations are done for d_(i)=“0” portions and when P^(3i) _(m) is “0”, so the selection of out-of-use bits determines the calculation quantity in case all of the 239 information bits are not 2o used. As the decoding includes a process of performing computation for searching the error position(s) after completion of the syndrome polynomial calculation, the calculation amount is desirably minimized in order to shorten a time taken for such calculation. This is attainable by performing selection of 128 optimal terms (degrees) from the 238-degree information polynomial f(x). This selection method will next be described below.

For the syndrome polynomials S₁(x) and S₃(x³), calculations are performed simultaneously in a parallel way. Calculation of each degree-term in each polynomial is the parity check of “1”; thus, the total calculation amount is expected to decrease if the coefficient of every degree is calculated without appreciable variations within almost the same length of time.

One preferred selection method therefore is arranged to include the steps of obtaining, for each “n”, a total sum of those with the coefficients being at “1” of these syndrome calculation-used 7-degree remainder polynomials p^(n)(x) and p^(3n)(x), and then selecting a specific number of n's corresponding to the required data bit number from the least side in number of the total sum. In this event, the first sixteen ones, i.e., the coefficients of x⁰ to x¹⁵, are used as the check bits to be fixed, and perform ascending-order selection of a total sum of “1”s of the coefficients to select 128 terms from the seventeenth one et seq.

Additionally, upon completion of the selection within a group of the same total-sum numbers, selection is done in order from the overlap of “1”s being less at the same degree terms as the reference while specifying n's as a reference with the coefficients “1” being uniformly distributed between respective degree terms within p^(n)(x) and p^(3n)(x) and then letting these n's be the reference. In other words, selection is done in order from the least side of the total sum of coefficients in the same terms as that of the reference with coefficients “1” of p^(n)(x), p^(3n)(x).

FIG. 11 shows 144 degrees “n” for use in the case of 144-bit data selected from 254 degrees in data polynomial f(x)x¹⁶ as described above.

Although this selection method does not always minimize the greatest one of the number of the coefficients “1” of respective degrees of the polynomial for execution of parity checking, it is still a simple method capable of reducing a step number of syndrome calculation while at the same time reducing the scale of syndrome calculator circuitry without requiring large-scale calculation including search-up of a calculation step-minimized one from among all possible combinations.

FIG. 12 shows a coefficient table of remainder polynomial r^(n)(x) obtained by g(x), i.e., a table of degree number “n”, at which the coefficient of remainder r^(n)(x) for selected x^(n) is “1”.

For example, the degree number “n” of r^(n)(x) with the coefficient of x¹⁵ being “1” is 17, 18, 22, . . . , 245, 249 and 250 written in fields defined by the number of coefficient “1” being from 1 to 62, in the column of m=15. Note that b₁₅ which is equivalent to the coefficient of a check bit x¹⁵ is obtainable as a result of parity check of this selected n-degree terms' coefficients in the information data polynomial f(x)x¹⁶.

FIG. 13 shows an exemplary circuit, which performs check bit calculation based on the table shown in FIG. 12. It is apparent from the table of FIG. 12 that for m=11, 5, 2 of x^(m), there are a maximum of 72 bit numbers to be subjected to parity check. So, this case is shown as an example in FIG. 13. Since there are indicated in the table is such the degree numbers “n” that the coefficient R^(n) _(m) of m-degree term of remainder polynomial r^(n)(x) is not “0”, select “n” from the table for each “m”, and then execute parity check using a_(n) or /a_(n).

An appropriate combination of parity check (PC) circuits to be used is determined depending on the number of inputs belonging to which one of the division remainder systems of four (4). More specifically, if it is dividable by 4, 4-bit PC is solely used. If such division results in presence of a remainder of 1, 5-bit PC is added. If the remainder is 2, 2-bit PC is added. If 3 remains then 3-bit PC is added.

In the example of m=11, 5 and 2, there are 72 inputs. A check bit calculator circuit adaptable for use in this case is configurable from three stages of parity check (PC) circuits as shown in FIG. 13. A primary stage consists of eighteen 4-bit PCs. The second stage is of 18 inputs, so let it be arranged by four 4-bit PCs and one 2-bit PC. The third stage becomes 5 inputs, so this is made up of a one 5-bit PC. An output of the third-stage parity check circuit becomes a check bit b_(m), /b_(m).

Similar calculation to that in the case of check bit calculation will be performed in the syndrome calculation, in a way set forth below.

FIG. 14 is a table of the number of degrees whose coefficient is “1” in 7-degree remainder polynomial p^(n)(x) for use in the calculation of the syndrome polynomial S₁(x). For example, the degree number n of p^(n)(x) with the coefficient of x⁷ being “1” is 7, 11, 12, . . . , 237, 242, 245 written in fields defined by the number of coefficient “1” being from 1 to 56, in the column of m=7. The coefficient of x⁷ of S₁(x) is obtained as a result of parity check of the coefficient of this selected n-degree term in the data polynomial v(x).

An example of electrical circuitry for performing calculation of the syndrome S₁(x) based on the table of FIG. 14 is shown in FIG. 15. It is apparent from FIG. 14 that in the case of m=6, 2, the number of bits to be parity-checked is 66 in maximum. This case is shown as an example in FIG. 15. As those degree numbers n with the coefficient P^(n) _(m) of m-degree term of remainder polynomial p^(n)(x) not being at “0” are listed in the table, select n from this table about each m and then execute parity check using data bits d_(n) and /d_(n).

A proper combination of parity checkers (PCs) used is determined depending on the number of inputs belonging to which one of the division remainder systems of 4. More precisely, if it is just dividable by 4, then 4-bit PC is solely used; if the division results in presence of a remainder 1, 5-bit PC is added; if the remainder is 2, 2-bit PC is added; and, if 3 remains then 3-bit PC is added.

In the example of m=6, 2, there are 66 inputs. So in this case also, three stages of parity check (PC) circuits are used to configure the intended calculator circuitry. The first stage is made up of sixteen 4-bit PCs and one 2-bit PC. The second stage is of 17 inputs, so it is constituted from three 4-bit PCs and a 5-bit PC. The third stage is of 4 inputs and thus is arranged by only one 4-bit PC. An output of the third stage becomes a coefficient (s1)_(m).

The same goes with the calculation of the syndrome polynomial S₃(x³), and this will be discussed below.

FIG. 16 is a table of the number of degrees whose coefficient is “1” in the remainder p^(3n)(x) for use in the calculation of syndrome polynomial S₃(x³).

For example, the degree number n of p^(3n)(x) with the m coefficient of x⁷ being “1” is 4, 8, 14, . . . , 241, 242 and 249 written in fields from 1 to 58 in the column of m=7. A data (s3)₇ that corresponds to the coefficient of x⁷ of S₃(x³) is obtainable as a result of parity check of the coefficient of this selected n-degree term in the data is polynomial v(x).

An exemplary calculation circuit therefor is shown in FIG. 17. As the FIG. 16 table suggests that the number of bits to be parity-checked for m=5 of x^(m) is 73 in maximum, this case is shown in FIG. 17 as an example. Since those degree numbers n with the coefficient P^(3n) _(m) of m-degree term of remainder polynomial p^(3n)(x) not being at “0” are shown in the table, select n from the table for each m and then execute parity check using d_(n) and /d_(n).

An appropriate combination of parity checkers (PCs) used is determinable depending on the number of inputs belonging to which one of the 4-division remainder systems. More specifically, if dividable by 4, use 4-bit PC only. If the division results in presence of a remainder 1, add 5-bit PC. If the remainder is 2, add 2-bit PC. If 3 remains then add 3-bit PC.

In the example of m=5, there are 73 inputs. Therefore, in this case also, three stages of parity check circuits are used to configure the calculation circuitry. The first stage is made up of seventeen 4-bit PCs and a 5-bit PC. As m the second stage is of eighteen inputs, it is formed of four 4-bit PCs and a 2-bit PC. The third stage may be configured from only one 5-bit PC as it becomes five inputs. An output of the third stage becomes the coefficient (s3)_(m), /(s3)_(m).

Next, it will be explained a method of making an error location searching circuit small in scale, which performs index comparison to search an error location(s).

A required calculation is to solve the index congruence shown in Expression 14. In detail, it is in need of solving two congruences. Firstly, obtain y_(n) of y²+y+1=α^(yn) based on the syndrome index. Then, find the index n of y=α^(n) corresponding to y_(n) based on the corresponding relationship, and solve the index i of x based on x=α^(σ1)y.

The congruences each is formed on GF(256), i.e., mod 255. If directly executing this calculation as it is, it becomes equivalent to performing the comparison of 255×255, and resulting in that the circuit scale becomes large. In this embodiment, to make the circuit scale small, the calculation will be performed in parallel.

That is, 2^(n)−1=255 is factorized into two prime factors M and N (i.e., first and second integers, respectively), and each congruence is divided into two congruences. Then, it will be used such a rule that in case a number satisfies simultaneously the divided congruences, it also satisfies the original congruence. In this case, to make the circuit scale and calculation time as small as possible when the congruence calculations are executed in parallel, it is preferred to make the difference between the two integers as small as possible.

In detail, 255 is dividable as 17×15, 51×5 or 85×3. In this embodiment, 17×15 is used, and two congruences will be solved simultaneously with mod 17 and mod 15.

First, to obtain y_(n), congruences shown in Expression 18 are used. That is, addition/subtraction with mod 17 between indexes multiplied by 15 and addition/subtraction with mod 15 between indexes multiplied by 17 are performed simultaneously in parallel.

15y_(n)≡15σ₃−45σ₁(mod 17)

17y_(n)≡17σ₃−51σ₁(mod 15)→y_(n)≡₃−3σ₁(mod 15·17)   [Exp. 18]

To obtain index i, the following Expression 19 is used. Here also, addition/subtraction with mod 17 between indexes multiplied by 15 and addition/subtraction with mod 15 between indexes multiplied by 17 are performed simultaneously in parallel.

15i≡15n+15σ₁(mod 17)

17i≡17n+17σ₁(mod 15)→i≡n+σ₁(mod 17·15)   [Exp. 19]

To execute addition and subtraction operations described above, an operation circuit is used, which is referred to as an index rotator described later. This circuit has a scale of product of numbers counting the different indexes with respect to mod, which are subjected to addition or subtraction. Therefore, if the above-described division of the congruence is not done, the calculation scale of the first congruence becomes 255×85=21675; and that of the second congruence 255×255=65025.

By contrast, by use of the above-described congruence division, the circuit scale of the first congruence becomes 17×17=289 plus 15×5=75, i.e., 364, which is a scale of about 1.7%. The circuit scale of the second congruence becomes 17×17=289 plus 15×15=225, i.e., 514, which is a scale of about 0.8%. As described above, the calculation scale is made small; and the calculation time shortened.

Next, the relationships between the respective indexes will be summarized below.

FIG. 18 shows the coefficient “1”, “0” at every degree number n of polynomial p^(n)(x), which is the reminder of division of x^(n) by m₁(x), and hexadecimal indications thereof of degrees 0 to 3 and 4 to 7. All index relationship explained below will be calculated based on this table. That is, each of four basic operations for the root α of m₁(x), which are performed by use of indexes, always corresponds to anywhere in this table with the relationship of one versus one.

FIGS. 19A and 19B show, with respect to the index range of n=0 to 255, index y_(n) of y²+y+1=α^(yn), coefficients of the corresponding polynomial and hexadecimal indications thereof. In case two “n”s correspond to the same y_(n), the number of errors is 2 while in case one “n” corresponds to one y_(n), there is one error. In case the number of errors is 2, it is shown that two indexes constituting a pair.

Note here that y²+y+1=0 at indexes 85 and 170. This means that S₃/S₁ ³ is 0 element of GF(256). In this case, there will be provided a control system, which directly controls an error correction circuit by use of the syndrome value.

FIG. 20 shows summarized relationships between index n and y_(n), in which two tables are arranged in parallel as follows: in one table, y_(n) is arranged in order of n; in the other, n is arranged in order of y_(n). In the latter table, it is shown that two “n”s correspond to the same y_(n) except y_(n)=0. At n=85 and n=170, there is no corresponding y_(n) (i.e., corresponding to element 0 of Galois field).

FIG. 21 shows that each index y_(n) is uniquely classified by 15 y _(n)(mod 17) and 17 y _(n)(mod 15). In the left half of FIG. 21, y_(n) is arranged from the least side of 15 y _(n)(mod 17) while in the right half, y_(n) is arranged from the least side of 17 y _(n)(mod 15).

It will be apparent from the table shown in FIG. 21 that congruence's paralleling as shown in Expression 18 leads to reduction of the calculation scale. The calculations for the congruences of 15 y _(n)(mod 17) and 17 y _(n)(mod 15) bring a common element, that becomes y_(n) obtained from the syndrome index.

FIG. 22 shows the relationship between indexes i selected as data bits and the corresponding physical bit locations k, and the index classification by 15 i(mod 17) and 17 i(mod 15). It is apparent from FIG. 22 that index i is classified into a pair defined by 15 i(mod 17) and 17 i(mod 15). It will be apparent from the table shown in FIG. 22 that congruence's paralleling as shown in Expression 19 leads to reduction of the calculation scale.

The calculations for the congruences of 15 i(mod 17) and 17 i(mod 15) bring a common element, that becomes i (i.e., k) obtained from index n and that of syndrome S₁ is when the physical bit location is calculated.

FIG. 23A shows an index rotator, which cyclically exchanges the passages from input nodes to output nodes to operate addition and subtraction of indexes; and FIG. 23B circuit symbol thereof. Since “subtraction” between indexes corresponds to the reversed shift in case of “addition”, addition will be explained below. Addition of indexes corresponds to product in the expression of root α. Therefore, two inputs of the rotator are expressed as α^(σi) and α^(σj); and output as α^(σi+σj).

As shown in FIG. 23A, the index rotator is for transferring inputs α⁰, α¹, α², . . . , α²⁵⁴ on the input nodes 201 to any of the output nodes 202 by use of only switch circuits 203 and wirings. Here is shown such a case that there are input and output nodes corresponding to all element on GF(256). In this case, input α⁰ may be transferred to any one of the output nodes 202 correspond to α⁰, α¹, α², . . . , α²⁵⁴ via the transistor switch circuits 203.

The gates of the switch circuit transistors serve as control input nodes 204, to which multipliers and divisors are input. That is, inverted signals of α⁰, α¹, α², . . . , α²⁵⁴ being used as control signals, a transistor corresponding to indexes to be added will be tuned on. As shown in FIG. 23A, in case N channel transistors are used as switching transistors, the input/output nodes are precharged, and the control signals controls which discharge passage becomes active between the input nodes and the output nodes.

Although, in the explanation described above, all element on GF(256) is used, element numbers relating to addition and subtraction are significantly different as dependent on what addition or subtraction is performed between indexes. In consideration of this, the solution method of a congruence is divided into two parts in parallel, thereby making the circuit scale small.

Next, it will be explained examples adapted practically to an error location searching part.

Index rotator 200 a shown in FIG. 24 is for operating the right side of the first congruence (i.e., 15 y _(n)≡15σ₃−45σ₁(mod 17)) in Expression 18. Input and control input are σ₃ and σ₁, respectively. Disposed on the input node side are 17 drivers 211 a, which decode coefficients (s3)_(m) (m=0 to 7) of 7-degree polynomial obtained by the syndrome calculation to drive an input signal at an index position of 15σ₃(17), i.e., the remainder class of 15σ₃ obtained by modulo 17.

Disposed on the control input side are 17 drivers 212 a, which decode coefficients (s1)_(m) (m=0 to 7) of 7-degree polynomial obtained by the syndrome calculation to drive a control input signal at an index position of −45σ₁(17), i.e., the remainder class of −45σ₁ obtained by modulo 17.

With this index rotator 200 a, an index corresponding to 15 y _(n)(17) in the table described above, i.e., the remainder class of 15 y _(n) obtained by modulo 17, is output to the output nodes.

FIG. 25 shows decoder circuits used in the drivers 211 a and 212 a shown in FIG. 24, each of which is formed of NAND circuits arranged in parallel in number of the irreducible remainder polynomials p^(n)(x) belonging to each remainder class. Each NAND circuit is formed of transistors connected in series with gates thereof being selectively applied with syndrome coefficients of m=0 to 7.

Precharge transistors are driven by clock CLK to precharge the corresponding common nodes. Whether the common nodes are discharged or not will serve as index signals of the remainder classes. Coefficient signal wirings and the inverted signal ones are disposed so as to constitute pairs, and these are selectively coupled to the gates of transistors in the NAND circuits in accordance with decoding codes.

The number of NAND nodes coupled in parallel is: 15 in case modulus of the remainder class is 17; and 17 in case that is 15.

Similar decode circuit is disposed on the control signal input node side, decoder code of which is shown in FIG. 26. In this table, indexes n of the irreducible remainder polynomial p^(n)(x) are classified into the remainder classes 15 n(17), which are obtained by modulo 17 with n being multiplied by 15. The remainders are classified by indexes 0 to 16, each of which includes 15 n's. In accordance with these coefficients of indexes of the corresponding p^(n)(x), it will be determined which signal wirings are coupled to decode transistor gates.

For example, in case of index 1, NAND nodes to be coupled in parallel are those of: n=161, 59, 246, 127, 42, 93, 178, 144, 212, 229, 110, 195, 8, 76 and 25.

FIG. 27 shows such a table that indexes n of the irreducible remainder polynomial p^(n)(x) are classified into the remainder classes −45 n(17), which are obtained by modulo 17 with n being multiplied by −45. The remainders are classified by indexes 0 to 16, each of which includes 15 n's. In accordance with these coefficients of indexes of the corresponding p^(n)(x), it will be determined which signal wirings are coupled to decode transistor gates.

For example, in case of index 1, NAND nodes to be coupled in parallel are those of: n=88, 173, 122, 156, 71, 20, 190, 207, 241, 54, 37, 139, 105, 224 and 3.

Index rotator 200 b shown in FIG. 24 is for operating the right side of the second congruence (i.e., 17 y _(n)≡17σ₃−51σ₁(mod 15)) in Expression 18. That is, this is for obtaining 17σ₃−51σ₁(mod 15) from indexes σ₃ and σ₁.

Inputs are σ₃. To decode coefficients (s3)_(m) (m=0 to 7) of 7-degree polynomial obtained by the syndrome calculation, and drive an input signal at an index position of 15σ₃(17), 15 drivers 211 b are disposed.

Control inputs are σ₁. To decode coefficients (s1)_(m) (m=0 to 7) of 7-degree polynomial obtained by the syndrome calculation, and drive a control input signal at an index is position of −51σ₁(15), 5 drivers 212 b are disposed. Since modulus is 15, 51 and 15 have a common prime 3. Therefore, remainder class number is 5 that is obtained by dividing 15 by 3, and the indexes of the remainder by modulo 15 are 0, 3, 6, 9, and 12. As a result, the number of drivers 212 b is 5; and that of control inputs is also 5.

Output to the output nodes are indexes, which designate the classes corresponding 17 y _(n)(15) in the above-shown table.

Decode circuits in the drivers are the same as shown in FIG. 24 except that input code is different from it.

FIG. 29 shows such a table that indexes n of the irreducible remainder polynomial p^(n)(x) are classified into the remainder classes 17 n(15), which are obtained by modulo 15 with n being multiplied by 17. The remainders are classified by indexes 0 to 14, each of which includes 17 n's. In accordance with these coefficients of indexes of the corresponding p^(n)(x), it will be determined which signal wirings are coupled to decode transistor gates.

For example, in case of index 1, NAND nodes to be coupled in parallel are those of: n=173, 233, 203, 23, 83, 158, 188, 68, 38, 128, 143, 98, 53, 218, 8, 113 and 248.

FIG. 30 shows such a table that indexes n of the irreducible remainder polynomial p^(n)(x) are classified into the remainder classes −51 n(15), which are obtained by modulo 15 with n being multiplied by −51. The remainders are classified by indexes 0, 3, 6, 9 and 12, each of which includes 51 n's. In accordance with these coefficients of indexes of the corresponding p^(n)(x), it will be determined which signal wirings are coupled to decode transistor gates.

For example, in case of index 3, NAND nodes to be coupled in parallel are those of: n=232, 22, 117, 122, 62, . . . , 47, 52, 27 and 2.

FIG. 31 shows bus structure, which integrates the operation results of the above-described two index rotators 200 a and 200 b to output operation signals to the following stage. To output the output signals of the rotators 200 a and 200 b, there are prepared output buses 215 a and 215 b, which are formed of 17 and 15 wirings, respectively.

With these buses 215 a and 215 b, 17+15 index data are transferred to the following decode part, which detects an error location(s) defined by the index of y. Here is shown that the operation result is inverted, and a selected index is output as a “H” signal.

Based on the signals on the output buses 215 a and 215 b, such decoding is performed as: from {15 y _(n)(17), 17 y _(n)(15)} to y_(n); from y_(n) to n; and from n to 15 n(17), 17 n(15), and the next state calculation is followed.

FIG. 32 shows index rotator 300 a, which operates the right side of the first congruence (i.e., 15 i≡15 n+15σ₁(mod 17)) in Expression 19. Inputs are signals on buses 215 a and 215 b. To decode these signals and input indexes of 15 n by modulo 17, 17 drivers 311 a are disposed.

Control inputs are σ₁. Disposed on the control input nodes 17 are 17 drivers 311 b, which decode coefficients (s1)_(m) (m=0 to 7) of 7-degree polynomial obtained by the syndrome calculation to drive a control input signal at an index position of 15σ₁(17), i.e., the remainder class of 15σ₁ obtained by modulo 17.

Obtained at the output nodes are indexes designating the remainder class of 15 i by modulo 17 corresponding to 15 i(17).

FIG. 33 shows decode circuits in the drivers 311 a disposed on the input side in FIG. 32. there are disposed a certain number of NAND circuits coupled in parallel with transistors connected in series, the number being determined in accordance with the relationships between the remainder classes. The gates of the NAND circuits are selectively applied with the signals corresponding to the remainder class indexes 17 yn(15) and 15 yn(17), which are output from the above-described rotators 200 a and 200 b. Each common node of NAND circuits connected in parallel, is precharged by precharge transistor driven by control signal /pr. Whether the common node is discharged or not serves as the index signal of a selected remainder class.

Although, in FIG. 33, some transistors in NAND circuit are shown as ones without gates, there are in practice no transistors at these positions, and only wirings are disposed. That is, practically connected in series in each NAND circuit are only two transistors.

Next, the relationships between remainder classes and codes serving as decoder inputs will be explained below.

FIG. 34 shows the relationships between the remainder class indexes 15 yn(17), 17 yn(15) and 15 n(17), and classes of yn and n corresponding thereto.

For example, {15 yn(17), 17 yn(15)} corresponding to the remainder class 15 n(17)=1 is as follows: {0, 7}, {1, 14}, {2, 13}, {4, 3}, {4, 4}, {7, 0}, {7, 11}, {8, 1}, {8, 14}, {11, 3}, {11, 13}, {12, 6}, {12, 12}, {15, 11} and {15, 14}. OR connections of these become decoder outputs.

That is, decoding is performed in accordance with this table, and outputs thereof become inputs of the index rotator 300 a.

FIG. 35 shows index rotator 300 b, which operates the right side of the second congruence (i.e., 17 i≡17 n+17σ₁(mod 15)) in Expression 19. Inputs are signals on buses 215 a and 215 b. To decode these signals and input indexes of 17 n in modulo 15, 15 drivers 311 b are disposed.

Control inputs are σ₁. Disposed on the control input nodes are 15 drivers 312 b, which decode coefficients (s1)_(m)(m=0 to 7) of 7-degree polynomial obtained by the syndrome calculation to drive a control input signal at an index position of 17σ₁(15), i.e., the remainder class of 17σ₁ obtained by modulo 15.

Obtained at the output nodes are indexes designating the remainder class of 17 i by modulo 15 corresponding to 17 i(15).

Decoder circuits used in the drivers are the same as shown in FIG. 33, and codes thereof are shown in FIG. 36. That is, FIG. 36 is a table showing the relationships between the remainder class indexes 15 y _(n)(17), 17 y _(n)(15) and 17 n(15), and classes of “y_(n)” and “n” corresponding thereto.

For example, {15 y _(n)(17), 17 y _(n)(15)} corresponding to the remainder class 17 n(15)=1 is as follows: {0, 10}, {1, 1}, {1, 13}, {2, 9}, {3, 1}, {4, 3}, {5, 12}, {6, 3}, {6, 8}, {11, 3}, {11, 8}, {12, 12}, {13, 3}, {14, 1}, {15, 9}, {16, 1} and {16, 13}. OR connections of these become decoder outputs.

That is, decoding is performed in accordance with this table, and outputs thereof become inputs of the index rotator 300 b.

FIG. 37 shows such a part that integrates the operation results of the index rotators 300 a and 300 b, and detecting error location y as a bit position. Outputs of the index rotators 300 a and 300 b are output to 15 i(17) bus 315 a and 17 i(15) bus 315 b, respectively.

It is apparent from the table shown in FIG. 22, which shows the relationship between “k”, “i”, 15 i(17) and 17 i(15), that “i” is uniquely designated by a pair of signals on these buses 315 a and 351 b. It is possible to specify “k” from the combination of {15 i(17), 17 i(15)}. Therefore, “k” will be selected via decoder circuits 316 each having two-input NOR gate. α^(i) becomes the final output of the operation result. One or two “k” is selected, and it designates error locations up to 2 bits.

FIG. 38 shows a summarized relationship between 15 i(17), 17 i(15), “i” and “k”, in which bit position indexes “i” are arranged in order of physical bit positions “k”, and remainder class indexes 15 i(17) and 17 i(15) also are shown as corresponding to the respective “i”.

A circuit for finally error-correcting based on the above-described operation result is shown in FIG. 39. The error correction circuit has different operations from each other in accordance with the syndrome operation results. In case S₁×S₃ is not 0, one or two errors have been generated, and error correction is performed. In case of S₁×S₃=0, there are two modes as follows: if s₁=s₃=0, there are no errors, and it is no need of error-correcting; and if one of S₁ and S₃ is 0, there are three or more errors, and it is not correctable.

To judge these situations, a judge circuit 401 with NOR gates G1 and G2 is prepared for detecting such a situation that all of the coefficients (s1)_(m) and (s3)_(m) of the syndrome polynomials is “0”. In case there are three or more error bits, one of the outputs of NOR gate G1 and G2 becomes “0”, and in response to it, NOR gate G5 outputs “1” that designates a non correctable state. In this case, NOR gate G4 outputs “0”, and this makes the decode circuit 402 performing error-correction inactive.

In case there are no errors, both outputs of NOR gates G1 and G2 are “1”, and NOR gates G4 and G5 also output “0”, thereby making the decode circuit 402 inactive.

If there is one bit error or two bit errors, both gates G1 and G2 output “0”, and output “1” of NOR gate G4 makes the decode circuit 402 with NOR gates G6 and G7 active. To invert dada d_(k) at the bit position k selected by α^(i), an inverting circuit 403 is prepared, which uses a two-bit parity check circuit shown in FIG. 40. With this inverting circuit 403, in case there are no errors, data d_(k) is output as it is while data d_(k) is inverted to be output at the error position.

According to this embodiment, the calculation scale and time of the ECC circuit with 2EC-BCH code will be effectively made small. That is, to perform error location searching in this embodiment, product and quotient operations of elements of GF(256) are executed as addition and subtraction operations with respect to indexes of the elements. In this case, 255 is divided into prime factors 15 and 17, and object numbers multiplied by 15 are subjected to addition/subtraction by modulo 17; other object numbers multiplied by 17 are subjected to addition/subtraction by modulo 15. These addition/subtraction operations are performed in parallel, and the final addition/subtraction result by mod 255 will be obtained from the results of two systems of the addition/subtraction operations.

The circuit scale of the index rotators for performing addition/subtraction of indexes will be reduced to about 2% or less in comparison with the case where the addition/subtraction circuit is formed as it is as performed by mod 255.

FIG. 41 shows a functional block configuration of ECC system 8 in the embodiment described above, in which memory core 10 is shown as one block. In encoding part 81, data m polynomial f(x)x¹⁶ based on the information polynomial f(x) is divided by code generation polynomial g(x)=m₁(x)m₃(x) to generate check bits, which are written into the memory core 10 together with to-be-written data.

Read out data from the memory core 10, which is expressed by v(x), are input to syndrome operation part 82 to be subjected to syndrome operations. In the error location searching part 83 for searching error location(s) based on the obtained syndrome coefficients, parallel operations are performed with index rotators 200 a and 200 b; and parallel operations for the operation results are further performed with index rotators 300 a and 300 b.

That is, index rotators 200 a and 200 b are for calculating the congruences shown in Expression 18, which compare index α₃−3σ₁ with index y_(n) obtained based on the variable conversion of: x=α^(σ1)y to find index y_(n) corresponding to an error location. Index rotators 300 a and 300 b are for calculating the congruences shown in Expression 19, which restore index yn to a real index “i” corresponding to the error location.

Error correcting part 84 inverts bit data at the error position.

This invention is applicable to any types of electrically erasable programmable semiconductor memory devices other than the flash memory as discussed in the above-noted embodiment.

FIG. 42 shows a configuration of one typical memory core circuit in a standard NAND flash memory, to which this invention is adaptable. This memory core circuit includes cell array 1 a, sense amplifier circuit 2 a and row decoder 3 a. Cell array 1 a is configured from parallel combination of NAND cell units (NAND strings) each having a serial connection of memory cells M0 to M31. NAND cell unit NU has one end connected to an associated bit line BLe (BLo) through a select gate transistor S1 and the other end coupled a common source line CELSRC via another select gate m transistor S2.

The memory cells have control gates, which are connected to word lines WL0-WL31. The select gate transistors S1-S2 have their gates coupled to select gate lines SGD and SGS, respectively. Word lines WL0-WL31 and select-gate lines SGD-SGS are driven by the row decoder 3 a.

The sense amp circuit 2 a has one page of sense units SA for performing “all-at-a-time” writing and reading. Each sense unit SA is associated with a bit line selector circuit 4, which selects either one of adjacent bit lines BLe, BLo for connection thereto. With such an arrangement, memory cells simultaneously selected by a single word line WLi and a plurality of even-numbered bit lines BLe (or odd-numbered bitlines BLo) constitute a page (one sector), which is subjected to a collective writing/reading. Bit w lines on the non-select side are used as shield lines with a prespecified potential being given thereto. This makes possible to suppress unwanted interference between the presently selected bit lines.

A group of NAND cell units sharing the word lines WL0-WL31 makes up a block, i.e., a unit for data erasure. As shown in FIG. 42, a predetermined number, n, of blocks BLK0-BLKn are laid out in the extending direction of the bit lines.

In the NAND flash memory with the core circuit also, the need grows for on-chip realization of correction of 2 bits or more errors with advances in the miniaturization and in per-cell multi-level storage scheme. The error correction system incorporating the principles of this invention is capable of reducing or minimizing the calculation scale for error detection and correction, thereby enabling successful achievement of computation at high speeds. Thus it can be said that the ECC architecture unique to the invention offers advantages upon application to memory chips of the type stated supra.

Since the above-described parallel operation method uses a general property with respect to the remainder class of the finite Galois field, it is not limited to 2EC-BCH system on GF(256), but is able to be adapted to other systems. For example, t(≧2)-error correcting BCH code may be used in general. In this case, it becomes such a BCH code on GF(2^(n)) that has t-element of α, α³, . . . , α^(2t−1) as roots thereof.

[Application Devices]

As an embodiment, an electric card using the non-volatile semiconductor memory devices according to the above-described embodiments of the present invention and an electric device using the card will be described bellow.

FIG. 43 shows an electric card according to this embodiment and an arrangement of an electric device using this card. This electric device is a digital still camera 101 as an example of portable electric devices. The electric card is a memory card 61 used as a recording medium of the digital still camera 101. The memory card 61 incorporates an IC package PK1 in which the non-volatile semiconductor memory device or the memory system according to the above-described embodiments is integrated or encapsulated.

The case of the digital still camera 101 accommodates a card slot 102 and a circuit board (not shown) connected to this card slot 102. The memory card 61 is detachably inserted in the card slot 102 of the digital still camera 101. When inserted in the slot 102, the memory card 61 is electrically connected to electric circuits of the circuit board.

If this electric card is a non-contact type IC card, it is electrically connected to the electric circuits on the circuit board by radio signals when inserted in or approached to the card slot 102.

FIG. 44 shows a basic arrangement of the digital still camera. Light from an object is converged by a lens 103 and input to an image pickup device 104. The image pickup device 104 is, for example, a CMOS sensor and photoelectrically converts the input light to output, for example, an analog signal. This analog signal is amplified by an analog amplifier (AMP), and converted into a digital signal by an A/D converter (A/D). The converted signal is input to a camera signal processing circuit 105 where the signal is subjected to automatic exposure control (AE), automatic white balance (AWB) control, color separation, and the like, and converted into a luminance signal and color difference signals.

To monitor the image, the output signal from the camera processing circuit 105 is input to a video signal processing circuit 106 and converted into a video signal. The system of the video signal is, e.g., NTSC (National Television System Committee). The video signal is input to a display 108 attached to the digital still camera 101 via a display signal processing circuit 107. The display 108 is, e.g., a liquid crystal monitor.

The video signal is supplied to a video output terminal 110 via a video driver 109. An image picked up by the digital still camera 101 can be output to an image apparatus such as a television set via the video output terminal 110. This allows the pickup image to be displayed on an image apparatus other than the display 108. A microcomputer 111 controls the image pickup device 104, analog amplifier (AMP), A/D converter (A/D), and camera signal processing circuit 105.

To capture an image, an operator presses an operation button such as a shutter button 112. In response to this, the microcomputer 111 controls a memory controller 113 to write the output signal from the camera signal processing circuit 105 into a video memory 114 as a flame image. The flame image written in the video memory 114 is compressed on the basis of a predetermined compression format by a compressing/stretching circuit 115. The compressed image is recorded, via a card interface 116, on the memory card 61 inserted in the card slot.

To reproduce a recorded image, an image recorded on the memory card 61 is read out via the card interface 116, stretched by the compressing/stretching circuit 115, and written into the video memory 114. The written image is input to the video signal processing circuit 106 and displayed on the display 108 or another image apparatus in the same manner as when image is monitored.

In this arrangement, mounted on the circuit board 100 are the card slot 102, image pickup device 104, analog amplifier (AMP), A/D converter (A/D), camera signal processing circuit 105, video signal processing circuit 106, display signal processing circuit 107, video driver 109, microcomputer 111, memory controller 113, video memory 114, compressing/stretching circuit 115, and card interface 116.

The card slot 102 need not be mounted on the circuit board 100, and can also be connected to the circuit board 100 by a connector cable or the like.

A power circuit 117 is also mounted on the circuit board 100. The power circuit 117 receives power from an external power source or battery and generates an internal power source voltage used inside the digital still camera 101. For example, a DC-DC converter can be used as the power circuit 117. The internal power source voltage is supplied to the respective circuits described above, and to a strobe 118 and the display 108.

As described above, the electric card according to this embodiment can be used in portable electric devices such as the digital still camera explained above. However, the electric card can also be used in various apparatus such as shown in FIGS. 45A to 45J, as well as in portable electric devices. That is, the electric card can also be used in a video camera shown in FIG. 45A, a television set shown in FIG. 45B, an audio apparatus shown in FIG. 45C, a game apparatus shown in FIG. 45D, an electric musical instrument shown in FIG. 45E, a cell phone shown in FIG. 45F, a personal computer shown in FIG. 45G, a personal digital assistant (PDA) shown in FIG. 45H, a voice recorder shown in FIG. 45I, and a PC card shown in FIG. 45J.

Additional advantages and modifications will readily occur to those skilled in the art. Therefore, the invention in its broader aspects is not limited to the specific details and representative embodiments shown and described herein. Accordingly, various modifications may be made without departing from the spirit or scope of the general inventive concept as defined by the appended claims and their equivalents. 

1. A semiconductor memory device comprising: a first cell array including a plurality of memory cells; a first sense amplifier circuit disposed adjacently to the first cell array on a positive column direction side of the first cell array and configured to read or write data of the memory cells in the first cell array; a second cell array disposed on a positive row direction side of the first cell array and including a plurality of memory cells; a second sense amplifier circuit disposed adjacently to the second cell array on the positive column direction side of the second cell array and configured to read or write data of the memory cells in the second cell array; and an ECC circuit disposed between the first cell array and the second cell array and configured to execute error processing of data read from the memory cells or written to the memory cells in the first cell array and the second cell array.
 2. The semiconductor memory device according to claim 1, further comprising: a first data bus extending in a row direction and configured to transmit or receive data between the first sense amplifier circuit and the ECC circuit; and a second data bus extending in the row direction and configured to transmit or receive data between the second sense amplifier circuit and the ECC circuit.
 3. The semiconductor memory device according to claim 1, further comprising: an input/output buffer configured to control input and output of data with external; a first data bus extending in a column direction and configured to transmit or receive data between the first sense amplifier circuit and the input/output buffer and between the ECC circuit and the input/output buffer; and a second data bus extending in the column direction and configured to transmit or receive data between the second sense amplifier circuit and the input/output buffer and between the ECC circuit and the input/output buffer.
 4. The semiconductor memory device according to claim 3, wherein the first data bus is disposed between the first cell array and the ECC circuit, and the second data bus is disposed between the second cell array and the ECC circuit.
 5. The semiconductor memory device according to claim 1, further comprising: a third cell array disposed on the positive column direction side of the first cell array so as to sandwich the first sense amplifier circuit and including a plurality of memory cells; and a fourth cell array disposed on the positive column direction side of the second cell array so as to sandwich the second sense amplifier circuit and including a plurality of memory cells.
 6. The semiconductor memory device according to claim 5, wherein the first sense amplifier circuit is further configured to read and write data of the memory cells in the third cell array, and the second sense amplifier circuit is further configured to read and write data of the memory cells in the fourth cell array.
 7. The semiconductor memory device according to claim 5, comprising: a first data bus disposed between the first cell array and the third cell array, and a second data bus disposed between the second cell array and the fourth cell array.
 8. The semiconductor memory device according to claim 1, wherein the first cell array and the second cell array each extend in the row direction and include a plurality of word lines for selecting the memory cells, and the semiconductor memory device further comprises: a first row decoder disposed adjacently to the first cell array on a negative row direction side of the first cell array and configured to select one of the word lines in the first cell array; and a second row decoder disposed adjacently to the second cell array on the positive row direction side of the second cell array and configured to select one of the word lines in the second cell array.
 9. A semiconductor device comprising: a first cell array including a plurality of memory cells; a first sense amplifier circuit disposed adjacently to the first cell array on a positive column direction side of the first cell array and configured to read or write data of the memory cells in the first cell array; a second cell array disposed on a positive row direction side of the first cell array and including a plurality of memory cells; a second sense amplifier circuit disposed adjacently to the second cell array on the positive column direction side of the second cell array and configured to read or write data of the memory cells in the second cell array; and an ECC circuit disposed between the first cell array and the second cell array and configured to execute error processing of data read from the memory cells or written to the memory cells in the first cell array and the second cell array, the first cell array and the second cell array each including: a plurality of bit lines extending in the column direction; a source line extending in the row direction; and a plurality of NAND strings, each of the NAND strings configured from a plurality of the memory cells connected in series, the memory cells being non-volatile and electrically rewritable, and a first and second select gate transistor for respectively connecting the two ends of the NAND string to one of the bit lines and the source line.
 10. The semiconductor memory device according to claim 9, further comprising: a first data bus extending in a row direction and configured to transmit or receive data between the first sense amplifier circuit and the ECC circuit; and a second data bus extending in the row direction and configured to transmit or receive data between the second sense amplifier and the ECC circuit.
 11. The semiconductor memory device according to claim 9, further comprising: an input/output buffer configured to control input and output of data with external; a first data bus extending in a column direction and configured to transmit or receive data between the first sense amplifier circuit and the input/output buffer between the ECC circuit and the input/output buffer; and a second data bus extending in the column direction and configured to transmit or receive data between the second sense amplifier circuit and the input/output buffer and between the ECC circuit and the input/output buffer.
 12. The semiconductor memory device according to claim 11, wherein the first data bus is disposed between the first cell array and the ECC circuit, and the second data bus is disposed between the second cell array and the ECC circuit.
 13. The semiconductor memory device according to claim 9, further comprising: a third cell array disposed on the positive column direction side of the first cell array so as to sandwich the first sense amplifier circuit and including a plurality of memory cells; and a fourth cell array disposed on the positive column direction side of the second cell array so as to sandwich the second sense amplifier circuit and including a plurality of memory cells.
 14. The semiconductor memory device according to claim 13, wherein the first sense amplifier circuit is further configured to read and write data of the memory cells in the third cell array; and the second sense amplifier circuit is further configured to read and write data of the memory cells in the fourth cell array.
 15. The semiconductor memory device according to claim 13, comprising: a first data bus disposed between the first cell array and the third cell array, and a second data bus disposed between the second cell array and the fourth cell array.
 16. The semiconductor device according to claim 9, wherein the first cell array and the second cell array each extend in the row direction and include a plurality of word lines for selecting the memory cells, and the semiconductor memory device further comprises: a first row decoder disposed adjacently to the first cell array on a negative row direction side of the first cell array and configured to select one of the word lines in the first cell array; and a second row decoder disposed adjacently to the second cell array on the positive row direction side of the second cell array and configured to select one of the word lines in the second cell array.
 17. A semiconductor memory device comprising: first through eighth cell arrays disposed in a column direction and a row direction and each including a plurality of memory cells; first through fourth sense amplifier circuits each configured to read or write data of the memory cells; and an ECC circuit configured to execute error processing of data read for the memory cells or written to the memory cells, the fifth cell array being disposed first in the row direction and first in the column direction; the sixth cell array being disposed fourth in the row direction and first in the column direction; the seventh cell array being disposed first in the row direction and second in the column direction; the eighth cell array being disposed fourth in the row direction and second in the column direction; the fifth through eighth cell arrays configuring a first memory bank; the first cell array being disposed second in the row direction and first in the column direction, the second cell array being disposed third in the row direction and first in the column direction, the third cell array being disposed second in the row direction and second in the column direction, the fourth cell array being disposed third in the row direction and second in the column direction, the first through fourth cell arrays configuring a second memory bank, the first sense amplifier circuit being disposed between the first cell array and the third cell array and being configured to read or write data of the memory cells in the first and third cell arrays, the second sense amplifier circuit being disposed between the second cell array and the fourth cell array and being configured to read or write data of the memory cells in the second and fourth cell arrays, the third sense amplifier circuit being disposed between the fifth cell array and the seventh cell array and being configured to read or write data of the memory cells in the fifth and seventh cell arrays, the fourth sense amplifier circuit being disposed between the sixth cell array and the eighth cell array and being configured to read or write data of the memory cells in the sixth and eighth cell arrays, and the ECC circuit being disposed between the first and third cell arrays and the second and fourth cell arrays.
 18. The semiconductor memory device according to claim 17, further comprising: a first data bus extending in the row direction and configured to transmit or receive data between the first and third sense amplifiers circuit and the ECC circuit; and a second data bus extending in the row direction and configured to transmit or receive data between the second and fourth sense amplifier circuits and the ECC circuit.
 19. The semiconductor memory device according to claim 17, further comprising: an input/output buffer configured to control input and output of data with external; a first bus data extending in the column direction and configured to transmit or receive data between the first and third sense amplifier circuits and the input/output buffer and between the ECC circuit and the input/output buffer; and a second data bus extending in the column direction and configured to transmit or receive data between the second and fourth sense amplifier circuits and the input/output buffer and between the ECC circuit and the input/output buffer.
 20. The semiconductor memory device according to claim 19, wherein the first data bus is disposed between the first and third cell arrays and the ECC circuit, and the second data bus is disposed between the second and fourth cell arrays and the ECC circuit. 